диафрагмированные волноводные фильтры / 75b4d4ea-900d-4059-9412-acd03a27d10c

Abstract — In this paper, a novel coupling structure for broad-band band-pass fin-line filter design is analyzed by 3-D full-wave simulation and then utilized as K inverter network. Network parameters are obtained and compared with those of the traditional inductive coupling structure, indicating that higher coupling strength can be achieved by choosing the proper size of the proposed coupling structure. Meanwhile, the physical length of the presented structure is bigger than its traditional counterpart. A 7-orders ka-band improved fin-line filter is designed and fabricated. In order to obtain broadband width and keep compact size simultaneously, both the proposed coupling structure and the traditional one are applied in the filter design. Measurements show that the fractional bandwidth of the filter is about 8%, and the insertion loss is less than 1.2 dB in pass-band of the filter. Moreover, the outmost coupling structure dimension is: w=12 mil, s=38 mil. There is no difficulty in physical realization.

Index Terms — Broad-band, coupling structure, fin-line, filter.

I. INTRODUCTION

E-plane filters have been reported in numerous literatures [1]-[7]. They are popular for their low cost, low loss and suitability for mass manufacturing techniques. E-plane fin– line band-pass filter usually consists of a ladder-type insert centered in the E-plane of rectangular waveguide, as shown in Fig. 1(a). The insert contains a piece of dielectric substrate to support a thin metallization pattern on one or both sides of the substrate and metallic strips of suitable width, which bridge the fin–line slot at distances of about half a wave– length, as shown in Fig. 1(b), thus performing inductive coupling between the transmission resonators. In order to minimize insertion loss the slot width g usually equals the waveguide height [5].

Owing to lacking effective design methods, many E-plane filter designs reported are limited to narrow bandwidths of less than a few percents. An accurate method for wide-band E-plane filter design was proposed by BUI et al. [6] in 1984. The method gave exact values for filter designs of up to 30 percent. However, for wideband filter design, as all the resonators are strongly coupled, especially the input and output coupling becomes very tight, the metallic strip of the traditional fin-line filter coupling structure becomes too narrow to physically realize. For example, provided the tolerable minimum value of the processing is 4 mil, the maximum available fractional bandwidth is about 9% for a 7- orders E-plane filter having 0.01dB pass-band ripple and 30 GHz center frequency with zero-thickness all-metal insert [7] and the maximum available bandwidth is still decreased as thickness of the insert is increased.

Waveguide

Dielectric

Substrate

(a)

g

Metallic strips

(b)

Fig. 1. (a) E-plane fin-line filter structure, (b) The insert configuration

Fig. 2. (a) The proposed coupling structure, (b) Side View

To effectively address this issue, this paper presents a novel fin-line coupling structure, which can realize tight coupling without increasing any processing difficulty. The proposed structure is analyzed and then utilized as K inverter for design broad-band fin-line filter. Based on it, a 7-orders ka-band modified fin-line filter using both the proposed coupling structure and traditional coupling structure is designed and optimized by using CST Design Environment. The simulation results agree well with the measurements.

II. EQUIVALENT CIRCUITS MODAL

The proposed coupling structure is realized by etching a gap in the middle of the metallic strip usually used in traditional fin-line filter coupling structure as demostrated in Fig. 2. Similar with the structure of variable height post in

(b)

Fig. 3. (a) Equivalent circuit of the proposed coupling structure, (b) Impedance inverter

Fig. 6. Characteristic impedance of impedance inverter K and electrical length versus gap width s

Fig. 4. The normalized reactance versus gap width s

Fig. 5. The normalized reactance versus metallic strip width w

rectangular wave-guide described in [8], the equivalent circuit of the proposed structure is modeled as a T-network as illustrated in Fig. 3(a). The normalized reactances Xp and Xs of the T-network can be obtained by the following expression [4]

Fig. 7. Characteristic impedance of impedance inverter K and electrical length versus metallic strip width w

where S11 and S12 are the scattering coefficients of the proposed coupled structure at the reference plane T. The S parameters can be calculated very accurately by using CST Design Environment. Since the normalized reactances Xs andXp in (1) are functions of the strip width w and width of the gap s, both parameters are plotted over w and s in Fig. 4 and Fig. 5 respectively. (All the curves are given at 30 GHz. The insert is realized by 10-mil-thick RT/duroid 5880 with the permittivity r=2.2 and using W-28 as a housing waveguide). As Fig. 4 illustrates, series resonant effects exhibited by the shunt reactance Xp are observed for a small gap width s and then Xp becomes more and more capacitive as s increases while keeping w constant. The polarity of the

Xp is mainly determined by s. It can be seen from Fig. 4 and Fig. 5 that the series reactance Xs is always inductive and weakly depends on s.

By connecting a transmission line in electrical length of/2 at both sides of the T-network, an impedance inverter is achieved, as shown in Fig. 3(b), where is determined by T- network parameters XS and XP.

The normalized characteristic impedance of K-inverter is obtained

Obviously, and K in (2) are also functions of the strip width and width of the gap. Fig. 6 and Fig. 7 demonstrate the relationships of and K versus w and s respectively (all the curves are given under the same conditions applied in Fig. 4 and Fig. 5). It can be seen from Fig. 6 that the characteristic impedance of K-inverter K is reduced rapidly as gap width s increases from zero to several mils (about 4 mil), then monotonically increased with gap width s. It is also observed that the maximum value of the K obtained by the proposed structure is significantly greater than its traditional counterpart (s=0 mil) with the same metallic width w. It means that the proposed structure based K-inverter can achieve higher coupling strength than its counterpart with the same metallic strip width w. In the meantime, electrical length of K-inverter in Fig. 3(b) is monotonically increased with the gap width s. In contrary, both K and are monotonically decreased with metallic strip width w, as depicted in Fig. 7. So a conclusion can be reached from Fig. 6 and Fig. 7 that a large electrical length is required for a high coupling strength K-inverter, therefore the physical length of the filter realized by the proposed structure is increased.

With the equivalent circuits discussed in the above paragraphs, it is indicated that the coupling strength is determined by width of the gap for a given metallic strip width. By choosing s and w value properly, a strongly coupling structure can be realized and broad-band fin-line filter can be achieved without increasing any processing difficulty.

III. SIMULATED AND MEASURED RESULTES

To verify the proposed design approach, a 7-orders finfilter is designed applying the procedure described in [6] and fabricated using a WR-28 waveguide and 10-mil-thick RT/duroid 5880 substrate. 30.2 GHz center frequency, 0.01 pass-band ripple and 8% fractional bandwidth are chosen for

Fig. 8. Photograph of the fin-line filter with the proposed coupling structure

Fig. 9. Measured and simulated frequency responses

the filter design. In order to miniaturize dimension of the filter, only the two outmost coupling elements are realized by the proposed structure, other coupling elements are realized by conventional metallic strip coupling structure. The filter is simulated and optimized by CST Design Environment. A photograph of the fabricated fin-line filter is shown in Fig. 8. The outmost coupling structure dimension is: w=12 mil, s=38 mil. There is no difficulty in physical realization. In order to obtain a clear comparison, the filter with the same parameters discussed above is also designed with all conventional metallic strip coupling structures and the two outmost metallic strips width is expected to be less than 1 mil. Obviously it can not be realized by adopting common processing.

Fig. 9 shows the simulated and measured results. It can be seen from the measured results that the insertion loss is less than 1.5 dB (include 0.3 dB insertion loss introduced by two back-to-back waveguide-to-SMA transitions used in measurement) and the return loss is more than 12 dB from 28.9 GHz to 31.3 GHz and the bandwidth is about 8%. As shown in Fig. 9, the measurements agree well with the simulations. The observed frequency deviation and the performance deterioration observed in pass band return loss are believed to be caused by processing mechanical error.

IV. CONCLUSION

This paper presented a new structure for broad-band finline filter design. In order to be compatible with conventional metallic strip coupling structure in fin-line filter design, an equivalent T-network has been given and analyzed then utilized as K inverter networks. Network parameters are studied and the results indicate that the proposed structure can achieve much tighter coupling than its counterpart and the physical length of it is also increased. Based on it, an improved fin-line filter adopting both the proposed coupling structure and the traditional one is designed and tested. The measured results have proved that the filter can achieve a broad bandwidth without increasing any processing difficulty. Moreover, the band width of the filter can be further increased by properly choosing the size of the proposed coupling structure.

ACKNOWLEDGEMENT

The author would like to thank Mr. Ye Tong for his CAD drawing and Mr. Weinan Wen for his support in filter measuring.

REFERENCES

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[2]Y. Konishi and K. Uenakada, “The design of a bandpass filter with inductive strip-planar circuit mounted in waveguide,”

IEEE. Trans. Microwave Theory Tech., vol.22, pp. 869-873, October 1974.

[3]F. Amdt, et al., “Theory and design of low-insertion loss finline filters,” IEEE. Trans. Microwave Theory Tech., vol.30, pp.155-162, February 1982.

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